Variable-Phase Ring-Oscillator Arrays, Architectures, and Related Methods

ABSTRACT

Embodiments of the present disclosure allow for a linear phase progression between adjacent elements in array by providing a symmetric ring configuration of tuned amplifiers and a single phase shifter. This ring topology is coupled to a single phase locked loop (“PLL”) that allows for direct modulation and demodulation of arbitrary waveforms without using RF up/down converting mixers. In addition, the PLL distributes the transmit waveforms to all antenna elements in the transmit mode and combines the received waveforms in the receive mode without any complicated power distribution network.

RELATED APPLICATION

This application claims priority to U.S. Provisional Application No.60/747,150 filed 12 May 2006, the content of which is incorporatedherein by reference in its entirety.

BACKGROUND

Phased arrays are spatially apart multiple antenna systems (a/k/aantenna arrays) that can focus the signal energy into a narrow beamradiating into/from specific directions and electronically change thedirection of signal transmission and reception. These beam formingschemes reduce the interference levels by placing a null in undesireddirections, increase the effective SNR, and conserve the battery powerby focusing the energy only at desired directions. Phased arrays havebeen used for radar, imaging, and communications in military, space,medical, and commercial applications due to their ability to formelectronically steerable beams. Phased arrays have been limited todiscrete implementations where physically separated antennas areconnected to their associated electronics (separate electronics for eachantenna). This approach leads to an increase in system cost, size, andpower consumption.

Phased arrays, also known as electronically steered arrays (ESA),imitate the behavior of directional antennas whose bearing can beadjusted electrically. They use multiple spaced antennas and can shapethe transmitted or received electromagnetic beam (beam forming). Inphased array receivers, the incident wave reaches spatially apartantenna elements at different times. This time delay difference is afunction of antenna spacing and angle of arrival. The receivercompensates for these time delays and combines all the signals toenhance the reception from the desired direction, while rejectingemissions from other directions (spatial selectivity). The coherentaddition of signal and the uncorrelated nature of noise in these systemsimprove the output signal-to-noise ratio by the array size.

Phased array transmitters provide appropriate time delay for the signalat each antenna element so that the radiated electromagnetic beam fromthe array has the intended shape (e.g., pointed to a specific angle).When a linear array of uniformly spaced antennas receives a planeelectromagnetic wave, each antenna receives a successively-time-delayedversion of the signal, with the inter-element delay depending on theinter-antenna separation and the angle of incidence of the wave. Thesignal can then be recovered with maximum power gain by compensating forthe inter-element delay electronically in the receiver. Consequently,exercising electronic control over the time-delay in each antenna'ssignal path allows one to “look” for electronic beams in differentdirections.

The design of any high performance integrated communication systembegins with the transceiver architecture. Phased arrays andtransmit/receive spatial diversity systems are the main applications ofmultiple antenna transceivers. Phased arrays imitate the behavior ofdirectional antennas whose bearing can be adjusted electrically. Theycompensate the time delay differences of the radiated signal between theantenna elements, and combine the signals to enhance the reception fromthe desired direction, while rejecting emissions from other directions.The coherent addition of signal and the uncorrelated nature of noise inthese systems improve the output SNR by the array size. Providing thedelayed version of the transmitted or received signal is a common uniquefeature of various multiple antenna schemes. In narrowbandimplementations, this time delay can be approximated with a constantphase shift. This approximation does not hold as the signal bandwidthgets larger and causes signal distortion. Controlling the signal timedelay in each path of a phased-array radio can be achieved by variousmethods involving multiple trade-offs in the performance of phased-arraysystems. Additionally, the amplitude of each path in multiple antennasystems can be individually controlled in order to increase the SNR andreduce the gain at undesired incident angles (controlling side-lobelevels and null location in the EM beam pattern).

FIG. 1 includes FIGS. 1(a)-1(b), which show two conventional approaches100 a and 100 b to phase shifting and power combining in the signal pathfor a phased array of antenna 106(1)-106(N) coupled to amplifiers102(1)-102(N). In narrowband systems such as shown in FIG. 1 (a(, themost straightforward method of adjusting the signal time delay is byproviding a variable phase shifter 104(1)-104(N) at the bandwidth ofinterest in each signal-path. The loss inevitable with most integratedvariable RF phase shifters reduces the receiver sensitivity and thetransmitter radiated power. Active implementations of RF phase shifterscan eliminate loss at the expense of increased nonlinearity and powerconsumption. However, by phase shifting and signal combining at RF, moreradio blocks are shared resulting in reduced area and power consumption.Additionally, since the unwanted interference signals are cancelledafter signal combining, e.g., by combiner 108, the dynamic-rangerequirements of the following blocks (both linearity and noise figure)such as mixer 110 coupled to local oscillator 112 and A/D 114, are morerelaxed. If amplitude control is needed, it can be achieved byvariable-gain low-noise amplifiers before or after the phase shifters atRF.

As depicted in FIG. 1(b), phase shifting and signal combining can alsobe performed after down-converting the received signals to anintermediate-frequency (IF). In FIG. 1(b), amplifiers 152(1)-152(N) arecoupled to antennas 156(1)-156(N). A mixer 158(1)-158(N) connected eachamplifier 152(1)-152(N) to a phase shifter 154(1)-154(N) after mixingwith a local oscillator 160. The phase shifter are connected to acombiner 162, which in turn is coupled to A/D 164.

With continued reference to FIG. 1(b), due to the additional signalamplification at the RF stages, phase shifter loss will have a lessdeteriorating effect on receiver sensitivity in case it is performed atthe IF stage, However, some of the aforementioned advantages, includinga lower dynamic-range requirement for the RF mixer, become lesseffective. Moreover, the value of passive components needed to provide acertain phase shift is inversely proportional to the carrier frequency.Since the value of integrated passive components is directly related totheir physical size, passive phase shifters at IF consume a larger area.

In wideband systems, a true variable time delay element should be placesin each signal path. For example, elements 104(1)-104(N) in FIG. 2(a)could be replaced with true variable time delay elements. The time delayof an EM wave can be varied by either changing the propagation velocity,altering length of signal propagation, or a combination of them.

FIG. 2 includes FIGS. 2(a)-2(b), which show two alternative conventionalapproaches 200 a and 200 b to phase shifting and power combining for aphased array. As shown in FIG. 2(a), phase shifting can be accomplishedin the local-oscillator (LO) path. The phase of the received signal canindirectly be varied by adjusting the phase of local-oscillator signalused to down-convert the signal to a lower frequency. This is due to thefact that the output phase of a multiplier (or mixer) is a linearcombination of its input phases. FIG. 2(a) shows a simplifiedphase-array receiver that uses LO phase shifting. Amplifiers202(1)-202(N), mixers 204(1)-204(N), antennas 206(1)-206(N), and phaseshifters 208(1)-208(N) are configured as shown. A local oscillator 210is used in conjunction with combiner 212 and A/D 214.

Phase shifting at the LO port is advantageous in that the phase shifterloss and nonlinearity does not directly deteriorate the receiver dynamicrange or transmitter radiated power. However, since the undesiredinterferences are only rejected after the combining step at the IF, RFamplifiers and mixers need to have a higher dynamic range compared tothe ones in the signal-path phase shifting scheme. The increased numberof building blocks might also increase the chip area and powerconsumption of the receiver. The control of signal amplitude can be madepossible more easily with IF variable-gain amplifiers (VGA). It shouldbe reminded that since the frequency of the local oscillator is fixed,the exact path delay can be maintained for only a single RF frequency.In other words, LO phase shifting is not an efficient solution forwide-band RF signals.

FIG. 2(b) depicts a digital array architecture used for a conventionalphased array application. The delay and amplitude of the received signalcan be adjusted at the baseband using a digital processor, as shown inFIG. 2(b). In FIG. 2(b), LO 250 is applied to mixers 252(1)-252(N)configured between amplifiers 250(1)-250(N) and A/D 256(1)-256(N).Amplifiers 250(1)-250(N) are connected to antennas 254(1)-254(N). DSP260 is connected to A/D 256(1)-256(N) as shown.

Digital array architectures such as shown in FIG. 2(b) can be veryflexible and can be adapted for other multiple antenna systems used forspatial diversity such as multiple-input multiple-output (MIMO) schemes.Despite its potential versatility, baseband phased-array architectureuses a larger number of components compared to the previous twoapproaches, resulting in a larger area, more power consumption, andhigher system complexity and cost. At the same time, since theinterference signals are not cancelled before baseband processing, allthe circuit blocks, including the power-hungry analog-to-digitalconverters (ADC), need to have a large dynamic range to accommodate allthe incoming signals without distortion. Above all, handling andprocessing a large amount of data through multiple parallel receiverscan be challenging even for today's advanced digital technology. Anillustrative example is shown in FIG. 2(b), where a baseband data-rateof 1.92 Gb/s is required. As a comparison, the fastest rate for sendingthe data into a personal computer using today's PCI standard is 32bits×33 MHz=1.056 Gb/s. This rate is almost halved when notebookcomputers are used (e.g., IEEE1394 standard supports 400 Mb/s).Alternatively, a very powerful digital signal processing (DSP) core canbe used to process this large influx of data, but it is going to bebulky, power-hungry, and expensive in today's technology.

In short, until faster and more power efficient digital data processingbecomes available at a lower price, digital implementations still is notan optimum solution for low-cost low-power multiple-antenna systems inhand-held applications (e.g. personal RADAR) or in sensory networks.

In addition to the aforementioned conventional approaches, coupledoscillator array have been utilized. In such, a linear array ofidentical second-order oscillators were each oscillator is coupled toits neighboring oscillators has a stable steady state response: alloscillators generate a sinusoid signal with the same frequency andphase. Upon manually controlling the phase of boundary oscillators, alloscillators will still generate the same frequency but with a linearphase progression from one end to the other end. In a linear narrowbandphased array, the EM plane wave radiates from adjacent antenna elementswith a linear phase difference. Therefore, coupled oscillator arrays areattractive solution to phased array implementations without usingexplicit phase shifters. The simple nearest neighbor coupling ofoscillators intended for a phased array application has seriousdrawbacks in practical implementations. Due to unavoidable mismatches inany implementation, the free running frequencies of integratedoscillators in an array are unequal and can vary by at least 1%. Thisseemingly small random frequency variation is sufficient to cause asignificant undesired phase shift. The desired phase shift betweenadjacent coupled oscillators have been set by (i) manually tuning thefree running frequency of each oscillator, or (ii) by having addingphase locked loops for all adjacent oscillator pairs. The first approachis not suitable for dynamic adjustment of phases for beam steering whilethe second approach adds to the system cost, complexity, and powerconsumption.

Phased arrays find wide use in radar, radio astronomy, remote sensing,electronic warfare, spectrum surveillance, wireless communications, andimaging applications. The advantage of phased array systems is morenoticeable as the number of elements in the array is increased. However,in conventional architectures, that translates to a significant increaseto the size, cost, and power consumption of the overall system. Phasedarray architectures that reduce the size and power consumption, whilepreserving the same performance, are highly desirable.

Higher frequencies offer more bandwidth for ultra high data ratewireless communications and better resolution for radar and imagingsystems, while reducing the required size of integrated systems in amulti-antenna configuration. Most of the existing phased array systemsare implemented using expensive processes and devices such as compoundsemiconductors. The reduction of minimum feature size in the metal oxidesemiconductor (MOS) transistor, accompanied by other rules of scalinghas resulted in a more dense integration and higher operation speed at alower cost for these circuits. Integration of a complete multi-antennasystem in a standard low-cost silicon process technology, especiallyCMOS, results in substantial improvements in cost, size, and reliabilityand provides numerous opportunities to perform on-chip signal processingand conditioning. Although newer silicon processing technologies offertransistors capable of operation at higher frequencies at a lower powerconsumption, other issues such as the reduction of power supply, the lowquality of integrated passive components, mismatch between integratedcomponents, and multiple sources of noise and interference propagatingin a conductive silicon substrate have hindered the proportionateadvancement of analog integrated communication circuits at extremelyhigh frequencies. Hence, system architectures and circuit techniquesthat allow for fully integrated silicon-based phased array solutions atradio-frequency (RF), microwave, and millimeter-waves are veryattractive.

In the recent past, the integration of phased array transceivers usingsilicon-based technologies has aroused a large amount of interest. Theseefforts are primarily motivated by economics, as silicon-basedtechnologies are far more cost-effective than more exotic technologies.However, the integration of phased array transmitters poses a number ofchallenges to the analog designer—a simple replication of the signalpath for each antenna results in significant area and power consumption.

SUMMARY

The present disclosure provides for a fundamentally different approachto beam-forming by integrating all the radiating elements (antennas) andelectronics on a single substrate, e.g., in a single standard siliconwafer. The radiated electromagnetic (EM) field from the silicon wafer iscontrolled by having several micro-radiators integrated in a closelyspaced mesh. These micro-radiators are controlled via integratedelectronics that operate at radio frequencies (RF), microwave, andmillimeter waves depending on the specific application, In fact, it willbe shown that the separation between radiating elements and the RFelectronics is very subtle in our proposed scheme: generation of thedesired RF beam and radiation occurs in one integrated architecture. Theco-design of radiators, RF electronics, and the signal processing coreall on the same silicon wafer results in substantial improvements incost, size, and reliability and opens up numerous possibilities in theDoD application space. The end result of the proposed work is a singlesilicon wafer connected to source of energy (i.e., battery) that canform arbitrary desired beam(s) for imaging, communication, and radar. Assuch, it can be used as a compact personal radar, communication device,or sensing element in a collaborative sensory network.

Embodiments of the present disclosure allow for a linear phaseprogression between adjacent elements in array by providing a symmetricring configuration of tuned amplifiers and a single phase shifter. Thisring topology is coupled to a single phase locked loop (“PLL”) thatallows for direct modulation and demodulation of arbitrary waveformswithout using RF up/down converting mixers. In addition, the PLLdistributes the transmit waveforms to all antenna elements in thetransmit mode and combines the received waveforms in the receive modewithout any complicated power distribution network.

When coupled with microwave circuit design techniques, such astransmission-line based design, the new architecture allows for theexploration of the upper frequency limits of standard digital processes.

Thus, embodiments of the present disclosure can allow for theelimination of a number of the building blocks traditionally seen inphased array communication systems, specifically delay-elements, mixersand RF power combiners. This is achieved through new blocks that performmore than one function, thus allowing for an integrated array that ismuch more compact and energy-prudent than its more traditionalcounterparts.

Systems, arrays, and architectures according to the present disclosurecan be used at and implemented for various desired frequency bands andapplications. In exemplary embodiments, such desired frequency bands andapplications can be GHz bands including the 22-29 GHz and 77-78 GHzbands for automotive radar applications, 24 GHz and 59-64 GHz IndustrialScientific Medical (ISM) bands for wireless local area networks, and the71-76 GHz, 81-86 GHz, and 92-95 GHz bands for point-to-point wirelesscommunications. Embodiments of the present disclosure may also implementand/or utilized other desired frequencies. For example, narrow bands ator around 2.4 GHz and 5 GHz may also be implemented.

BRIEF DESCRIPTION OF THE DRAWINGS

Aspects of the disclosure may be more fully understood from thefollowing description when read together with the accompanying drawings,which are to be regarded as illustrative in nature, and not as limiting.The drawings are not necessarily to scale, emphasis instead being placedon the principles of the disclosure. In the drawings:

FIG. 1 includes FIGS. 1(a)-1(b), which depicts two conventionalapproaches 100 a and 100 b to phase shifting and power combining in thesignal path for a phased array;

FIG. 2 includes FIGS. 2(a)-2(b), which depict two alternativeconventional approaches 200 a and 200 b to phase shifting and powercombining for a phased array;

FIG. 3 includes FIGS. 3(a)-3(b), which depict alternative embodiments300 a-300 b of a one-dimensional variable phase ring oscillatoraccording to the present disclosure;

FIG. 4 includes FIGS. 4(a)-4(d), which depict a one-dimensional variablephase ring oscillator coupled to a phase-locked loop according to anembodiment, as well as the magnitude and phase of second-order transferfunctions associated with a representative tuned oscillator;

FIG. 5 depicts an embodiment of a variable-phase ring oscillator coupledto antennas in a one-dimensional array operating in receive mode;

FIG. 6 depicts an embodiment of a two-dimensional m×n arrayarchitecture;

FIG. 7 depicts an embodiment of a multi-frequency beam forming arrayaccording to the present disclosure, with one-dimension shown forsimplicity;

FIG. 8 depicts a two-element embodiment as constructed on a die with0.13 micron CMOS technology and implementation at 70 GHz;

FIG. 9 depicts a method of applying a signal to a variable-phase ringoscillator architecture, in accordance with exemplary embodiments of thepresent disclosure.

FIG. 10 is a photograph with circuit overlay of a 24 GHz 0.13 micronCMOS variable phase ring oscillator transmitter and receiverarchitecture according to an exemplary embodiment of the presentdisclosure; and

FIG. 11 is an enlargement of a portion of FIG. 10 along with acorresponding circuit diagram showing

While certain embodiments are shown in the drawings, one skilled in theart will appreciate that the embodiments depicted in the drawings areillustrative and that variations of those shown, as well as otherembodiments described herein, may be envisioned and practiced within thescope of the present disclosure.

DETAILED DESCRIPTION

Embodiments of the present disclosure are directed to variable-phasering oscillator arrays, architectures, and related methods that allowfor a linear phase progression between adjacent elements in array byproviding a symmetric ring configuration of tuned amplifiers and asingle phase shifter. Such ring topologies can be coupled to a singlephase locked loop (“PLL”) that allows for direct modulation anddemodulation of arbitrary waveforms without using RF up/down convertingmixers. In addition, the PLL distributes the transmit waveforms to allantenna elements in the transmit mode and combines the receivedwaveforms in the receive mode without any complicated power distributionnetwork. Additional PLLs may be used for concurrent beam forming.

Exemplary embodiments of the present disclosure provide for afundamentally different approach to beam-forming by integrating all theradiating elements (antennas) and electronics on a single substrate,e.g., in a single standard silicon wafer. The radiated electromagnetic(EM) field from the silicon wafer is controlled by having severalmicro-radiators integrated in a closely spaced mesh. Thesemicro-radiators are controlled via integrated electronics that operateat radio frequencies (RF), microwave, and millimeter waves depending onthe specific application. Generation of the desired RF beam andradiation can occur in one integrated architecture. The co-design ofradiators, RF electronics, and the signal processing core all on thesame silicon wafer results in substantial improvements in cost, size,and reliability. Accordingly, exemplary embodiments can be implementedon a single semiconductor (e.g., silicon) wafer and can be connected tosource of energy (i.e., battery) that can form arbitrary desired beam(s)for imaging, communication, and/or radar. As such, it can be used as acompact personal radar, communication device, or sensing element in acollaborative sensory network.

To form arbitrary beam pattern(s), the EM field distribution on awafer/substrate, e.g., a silicon wafer can be actively controlled. Forsuch control, the amplitude and phase of the RF signal in each antennasite, e.g., μ-site, on the wafer/substrate, can be appropriatelycontrolled according to techniques of the present disclosure. For,example, at the center of each antenna site, a μ-radiator can be locatedthat can efficiently couple the generated EM wave in to air. Appropriatecircuitry, e.g., as shown and described herein, may be coupled to theantenna site(s) to generate a modulated waveform in transmit mode andrecover the information signal from the RF carrier in receive mode.

Embodiments of the present disclosure can provide for one or more of thefollowing, alone or in any combination: (i) One-dimensional arrayarchitecture for single frequency beam-forming; (ii) Two-dimensionalarchitecture for single frequency beam-forming; (iii) Concurrentmulti-frequency beam-forming arrays; and (iv) RF, microwave, andmillimeter wave circuit design and silicon integration. A standardlow-cost silicon process technology that can be used, which allows forthe integration of high frequency analog front-end and digital signalprocessor (DSP) on the same substrate or package. Such architectures canreduce the size, cost, and power consumption compared to existing andbrute force approaches, e.g., allowing techniques of the presentdisclosure to be used in wide deployment in mainstream cars andautomotive application. Other advantages and benefits are also withinthe scope of the present disclosure.

FIG. 3 includes FIGS. 3(a)-3(b), which depict alternative embodiments300 a-300 b of a one-dimensional variable phase ring oscillatoraccording to the present disclosure. FIG. 3(a) shows a one-dimensionalvariable phase ring oscillator architecture 300 a suitable for singlefrequency beam-forming. The architecture 300 a includes a ringconfiguration of identical second-order nonlinear elements, e.g., suchas LC-tunable (or tuned) amplifiers 302(1)-302(N), and a delaystructure, e.g., a narrowband phase shifter 306. The tuned amplified caninclude an inductor and variable capacitor (or varactor) in a tankcircuit, e.g., 304(1)-304(N), as shown.

The architecture 300 a shown in FIG. 3(a) is capable of generating asinusoidal waveform if the loop gain is larger than one. In this case,the output voltage of all tunable amplifiers 302(1)-302(N) is a sinusoidat the same frequency. The phase shift between adjacent node voltages isequal to a value that satisfies the phase boundary condition around theloop. Therefore, a linear phase progression, Δφ, between adjacentelements 302(1)-302(N) is realized and it can be continuously varied viaa single phase shifter based on the following expression:$\begin{matrix}{{\Delta\phi} = \frac{{2k\quad\pi} - \varphi}{N}} & (1)\end{matrix}$where φ is the phase shift provided by phase shifter, N is the number ofelements in the ring configuration, and k is any arbitrary integer. Forinstance, in the absence of phase shifter, one steady-state solution iswhere all nodes produce sinusoidal oscillations with the same phase assymmetry dictates. As the phase shift between adjacent elements (inputand output of tunable amplifiers) varies the oscillation frequencychanges as well. This can be understood by noticing the phase transferfunction of a second-order resonator as shown in FIG. 3(a). There aremany phase shifts, and hence many oscillation frequencies, that satisfyexpression (1) depending on the value of k. The present inventors haveshowed that all these modes are stable. As further shown and describedherein, one method to fix or control this frequency by using aphase-locked loop (PLL).

With continued reference to FIG. 3(a), ring configuration of 300 a canproduce the linear phase progression of RF signal necessary for linearphased arrays (i.e., each node is driving an antenna element). Tocontrol the phase and amplitude of RF signal in an arbitrary way inorder to set any desired EM field distribution on the silicon wafer, atunable resonant load is used, as shown 304(1)-304(N). As the resonantfrequency and hence the input-output phase and magnitude transferfunction of the tuned amplifiers is a function of its capacitor value,for such tuning, an integrated variable capacitor or varactor can beused. In addition, the amplitude of signal at each node can also be setindividually by applying appropriate bias to each amplifier. According,desired arbitrary signal phase and magnitude can be provided at eachnode by applying appropriate control voltages to the varactors load ofeach tuned amplifier as long as the loop boundary conditions aresatisfied.

FIG. 3(b) shows another embodiment or configuration of a one-dimensionalvariable phase ring oscillator architecture 300 b suitable for singlefrequency beam-forming. The architecture 300 b includes a ringconfiguration of identical second-order nonlinear elements, e.g., suchas amplifier differential pairs 352(1)-352(N), and a delay structure,e.g., a phase shifter 354.

With continued reference to FIG. 3(b), one configuration of a suitableamplifier differential pair 352(4) is shown enlarged. For such, inductor360, resistor 362, coupled transistors 364(1)-364(2), 366(1)-366(2), andbias current source 368 are shown.

In beam-forming applications, the frequency of operation and beamform/angle should be controlled independently. In order to form adifferent beam, the phase shift and amplitude of each radiator isvaried, however, that should not come at the expense of shifts infrequency. In order to fix the operating frequency in the architecturesaccording to the present disclosure, a ring confirmation can be coupledto a PLL as shown in FIG. 4.

FIG. 4 includes FIGS. 4(a)-4(d), which depict a one-dimensional variablephase ring oscillator architecture 400 a according to an embodiment, aswell as the magnitude and phase of second-order transfer functionsassociated with a representative tuned oscillator. In FIG. 4(a),architecture 400 a includes a series of tuned amplifiers 402(1)-402(N)in a ring with phase shifter 404 that is coupled to a phase-locked loop(PLL) 405. PLL 405 includes frequency divider 406 connected to frequencyphase detector (PFD) 408 connected to charge pump (CP) 410 in turnconnected to loop filter 412. Control voltage (V_(cntrl)) is suppliedfrom the PLL 405 to the tuned amplifiers 402(1)-402(N), as shown. InFIG. 4(a), two feedback loops work in conjunction: the phase loop setsthe necessary phase shift between adjacent elements in the ringconfiguration while the frequency loop forces the oscillation frequencyto be equal to that of a reference.

The PLL shown also has two other important functions: it can serve tomodulate arbitrary transmit waveforms and distribute them to allelements without any complicated power distribution network ormodulating mixers that are common in traditional schemes; the PLL cabalso demodulate arbitrary received waveforms and combine them withoutusing down converters, RF or LO signal distribution networks, or powercombiners as will be explained supra.

FIG. 4(b) depicts the representative tunable oscillator 402 as utilizedfor FIG. 4(a). FIG. 4(c)-4(d) depict the magnitude and phase ofsecond-order transfer functions, respectively, associated with therepresentative tunable 402 oscillator of FIG. 4(b);

FIG. 5 depicts an embodiment of a variable-phase ring oscillator coupledto antennas in a one-dimensional array 500 operating in receive mode.The architecture shown in FIG. 5 is similar to the ring architecture 400a with PLL 405 combination that was described for FIG. 4(a), with theaddition that each node of the ring is connected to an antenna (e.g.,μ-radiator) 504(1)-504(N). Phase shifter 506, ids shown connected to aPLL including frequency divider 518 connected to frequency phasedetector (PFD) 510 connected to charge pump (CP) 512 in turn connectedto loop filter 514. Control voltage (V_(cntrl)) is supplied from the PLLto the tuned amplifiers 502(1)-502(N), as shown.

With continued reference to FIG. 5, in operation of the array 500, theinformation signal at baseband is added to the varactors control voltageinside the PLL. In the absence of this signal, the control voltage valuewill settle to a constant value set by PLL in order to force thefrequency of the ring structure equal to a multiple of referencefrequency, ω_(ref). Each antenna will then radiate a different phase ofthe same frequency. The relative phase of the signal at each nodedepends on the value of variable phase shifter, number of tunedamplifiers, and the value of their variable resonant frequency. Intransmit mode, a baseband signal can be explicitly added (not shown) tothe control voltage node, which allows the PLL to force the frequency ofring structure to vary correspondingly.

In other words, at steady state the ring structure generates a modulatedwaveform around center frequency ω_(ref). The phase of radiated signalfrom each node at the RF frequency for the modulated waveform is stilldictated by the ring structure. The control voltage that is applied toeach element and the amplifier bias current can be varied independentlyin order to achieve arbitrary phase and amplitude at each node. This isuseful to control the levels of sidelobes, location of nulls, number ofmain beams, and their beam widths at a single carrier frequency. Directmodulation of PLL eliminates RF up-converters, RF power splitters, andLO distribution networks that are necessary in conventional schemes.

With continued reference to FIG. 5, topology or architecture 500 canalso demodulate the incident wave without adding extra components, asindicated. First, let us consider the ring architecture of interestwithout the PLL when exposed to a plane wave at ω_(inc) incident from anangle. Depending on the closeness of injected frequency to ring'snatural frequency, the strength of incidence wave, and incidence angletwo scenarios can happen. The first case is where the naturaloscillation frequency of the ring changes and locks to ω_(inc). Lockingrange, Δω_(lock), the difference between ring's natural frequency andincidence frequency where locking happens, can be calculated to be$\begin{matrix}{{\Delta\omega}_{lock}\frac{\omega_{0}{ɛ( {1 + {\tan^{2}({\Delta\phi})}} )}^{\sin}( \frac{N( {{\Delta\quad\theta} - {\Delta\phi}} )}{\quad} )}{{2Q} + {{\tan({\Delta\phi})}{N_{\sin}( \frac{{\Delta\theta} - {\Delta\phi}}{2} )}}}} & (2)\end{matrix}$

where Q is the quality factor of each resonator, ω_(o) is the ringoperating frequency, ε is the relative strength of incident wavecompared to the natural signal at each node of the ring, and α is theincidence angle. Expression (2) shows that Δω_(lock) contains the beamsteering information (array factor) as a function of incidence angle.Phase distribution across the ring under locked condition derives from$\begin{matrix}{{\tan^{- 1}( {Q( {1 - \frac{\omega^{2}}{\omega_{c}^{2}}} )} )} = \frac{{2k\quad\pi} + {\Delta\phi}}{N}} & (3)\end{matrix}$

where k is any arbitrary integer. Depending on the value of k there arevarious regions where locking can happen. In each region, the lockingrange is a function of incidence wave angle as predicted by (2). If theincidence frequency, ω_(inc), is outside the locking range, the ringfrequency gets pulled towards ω_(inc) and side tones appear around themain frequency.

With continued reference to FIG. 5, in the presence of PLL, the controlvoltage, V_(cntrl), that adjusts the center frequency of each resonatoris given by $\begin{matrix}{V_{cinrl} = {\frac{{\Delta\omega}_{lock}}{K_{vca}}{\sin( {{{\Delta\omega}_{inc}t} + {\theta_{BB}(t)}} )}}} & (4)\end{matrix}$

where K_(vco), often referred to as VCO gain, measures the sensitivityof ring frequency to the changes in the control voltage and Δω_(inc) isthe instantaneous difference between the frequency of incident wave andoscillators' natural frequency. In conclusion, the amplitude ofV_(cntrl) has the beam steering information (incidence angle, peaks,nulls) while the frequency of V_(cntrl) contains the down-convertedbaseband information, θ_(BB)(t). It is important to note that thecombination of the proposed ring array architecture and the PLL derivesthe beam steering information and modulated signal without using anypower splitters, any RF mixers, and N phase shifters.

Embodiments of the present disclosure provide the ability to arbitrarilyform the EM beam(s) using large number of closely spaced radiatingelements on a silicon wafer. As noted previously, one of the majorlimitations of conventional schemes such as the coupled oscillator arrayis their unacceptable sensitivity to mismatches in the array.Embodiments of the present disclosure are more robust and are resistantto such mismatches, e.g., by an order of magnitude compared to thecoupled oscillator array. As an example, the error in the beam pointingangle in exemplary embodiments according to the present disclosure,θ_(error), can be represented by $\begin{matrix}{\theta_{error} = {\frac{2Q}{n\quad{\pi cos}\quad{\theta_{0}( {1 + {\tan\quad{\Delta\phi}}} )}}{\sum\limits_{i = 1}^{n}{( {1 - \frac{6( {i - 1} )( {n - i + 1} )}{n^{2} - 1}} )( \frac{{\Delta\omega}_{i}}{\omega_{0}} )}}}} & (5)\end{matrix}$

where θ_(o) is the beam pointing angle with no mismatches, ω_(o) is thenominal resonance frequency of elements with no mismatches, and Δω_(i);is the frequency deviation of each resonator from the nominal value dueto component mismatches. The beam pointing accuracy improves with numberof array elements, N, and the beam pointing angle, θ_(o).

One-dimensional beam-former architecture can be easily extended totwo-dimensional (2D) arrays. FIG. 6 depicts an embodiment of atwo-dimensional m×n array architecture 600 of antenna (e.g., 604_(1,1),etc.). Phase shifter for each dimension of the array are shown,606(1)-606(N) and 608(1)-608(M). Transmit waveform generator 610 isshown connected to PLL 608 which is connected to the array and isconfigured to produce down-converted received wave form 612.

In a 2D architecture, such as shown in FIG. 6, each tuned blockamplifies the summation of signals from its adjacent row and column asshown in FIG. 6. By using such a scheme, electromagnetic beam(s) can bepointed to arbitrary elevation and azimuth. In conventional 2D N×N (orN×M) array architectures, a separate phase shifter per antenna elementis required. In our proposed scheme, only one shifter per row and percolumn of the array is required and hence the total number of phaseshifters is reduced from N² to 2N (or N×M to N+M).

One of the features provided by the present disclosure is a fullyintegrated architecture where multiple independent beams at differentfrequencies can be formed and electronically scanned towards independenttargets, simultaneously. Each of these independent beams can be used forcommunication, sensing, imaging, or other purposes, e.g., radarapplications. FIG. 7 depicts one such embodiment of a multi-frequencybeam forming array 700 according to the present disclosure, withone-dimension shown for simplicity.

In FIG. 7, the multi-frequency beam forming array 700 includes antennas702(1)-702(N) configured for a first frequency ω1 and antennas704(1)-704(N) configured for a second frequency ω2. Tuned amplifiers706(1)-706(N) include doubly resonant (fourth-order) circuits708(1)-708(N). Array 710 is depicted by a concurrent dual-frequency ringstructure with a fourth-order PLL 711. A single phase shifter 710 withtwo degrees of freedom modifies the phase of its input signal at eachfrequency by an independent amount. PLL include frequency divider 712coupled to phase detectors 714(1)-714(2) and loop filters 716(1)-716(2)for producing control voltages Vcntrl,1, Vcntrl,2 based on thereferences shown.

In steady-state, each node of the ring of 700 generates two frequencies,while a single phase shifter adjusts the phase progression betweenadjacent elements at both frequencies, independent of each other.Therefore, arbitrary and independent beams can be formed at twofrequencies, concurrently. Modulation and demodulation can be doneinside the fourth-order PLL similar to the single frequency case, e.g.,described for FIG. 3.

Accordingly, embodiments similar to that of FIG. 7 offer advantages overthe prior art as using conventional approaches discussed previously, alarge number of independent RF transceivers, phase-shifters, powercombiners, and ADCs are required to achieve concurrent multi-frequencybeam-forming. Specifically, if concurrent operation at m frequency bandsis desired, and the number of antenna elements in a one-dimensionalarray is N, a total of m×n RF transceivers are required in conventionalapproaches.

It should be understood that although the schematic is shown in FIG. 7for the case of one-dimensional concurrent dual-frequency beam-former,the idea can be extended to support more concurrent frequencies (byusing higher order resonators for each amplifier), and totwo-dimensional arrays; the nonlinear analysis of sixth- andhigher-order systems is more involved.

The beam-forming ring architecture of FIG. 7 includes higher ordermulti-resonant systems 706 (1)-706(N)—replacing the second-order systemsshown in FIG. 3. Such a system has multiple stable modes of operation insteady-state, specifically, it can generate stable oscillations at anyof resonant frequencies and more interestingly it can generate stableoscillations at all resonant frequencies, simultaneously. It should beemphasized that in general the resonant frequencies are completelyindependent of each other and therefore the output, which is a sum ofindependent sinusoids, does not have to be a periodic function(asynchronous oscillations). The multi-resonant ring is a special caseof multi-mode oscillators.

Using nonlinear analysis in the case of a fourth-order system(dual-frequency oscillator), we were able to find and implement methodsthat control the steady-state stable mode: oscillations at ω_(t),oscillations at ω₂, or simultaneous oscillations at ω₁, and ω₂.Moreover, we have recently showed that a not only a single oscillatorcan generate simultaneous frequencies, by also can vary each frequencyindependently using a single high-order phase-locked loop.

FIG. 8 depicts a two-element embodiment 800 as constructed on a die with0.13 micron CMOS technology and implementation at 70 GHz. As shown, fourtuned amplifiers 802(1)-802(4) were constructed in a ring topology,connecting two ring antenna 804(1)-804(2) by way of LNA/PA808(1)-808(6). For the embodiment, the PA output power was 11 dBm,antenna efficiency was 11%, and the die area was 3.2 mm×1.3 mm.

FIG. 9 depicts a method of 900 applying a signal to a variable-phasering oscillator architecture, in accordance with exemplary embodimentsof the present disclosure. A signal can be applied to a variable phasering oscillator coupled to a plurality of antennas, as described at 902.An integer number of successive phase delays can be provided to thesignal with the ring oscillator, as described at 904. The frequency ofthe ring oscillator can be locked with a phase delay structure, asdescribed at 906. A phase shift and amplitude of a signal can beassociated with a direction for each of the plurality of antennas, asdescribed at 908.

Exemplary embodiments directed to concurrent beam-forming can includeapplication of a signal of a second (or additional) frequency(cies) tothe plurality of antennas coupled to the variable phase ring oscillator,as described at 910.

With continued reference to FIG. 9, the method cab include applying asignal to a variable phase ring oscillator that include transmitting asignal from variable phase ring oscillator coupled to the plurality ofantennas. Applying a signal to a variable phase ring oscillator caninclude receiving a signal from variable phase ring oscillator coupledto the plurality of antennas. The method 900 for transmittingapplications can include modulating a carrier signal, which can includemodulating a carrier signal comprises a phase modulation scheme. Themodulation scheme comprises BPSK modulation.

Receiving according to method 900 can include demodulating a carriersignal. Demodulating a carrier signal can include a phase demodulationscheme, which can in certain embodiments be selected from the groupconsisting of QAM, OFDM, BPSK, AM, FM, PM, QPSK.

Embodiments of method 900 can include determining a distance to a targetobject using the signal applied to the variable phase ring oscillatorcoupled to a plurality of antennas. Method 900 can include applicationof a GHz signal. In exemplary embodiments, such can be a 22-29 GHz,59-64 GHz, 71-76 GHz, 77-78 GHz, 81-86 GHz, or 92-95 GHz band.

Further embodiments according to FIG. 9, can include that determining adistance comprises determining a distance from an object to anautomobile connected to the variable phase ring oscillator coupled tothe plurality of antennas.

Exemplary Embodiments RF, Microwave, and Millimeter Wave Circuit Designand Silicon Integration

Design and implementation of RF, microwave, and millimeter wave circuitsusing a silicon technology (CMOS and SiGe) is one of the keyenablers/factors in the realization of embodiments of the presentdisclosure. The present inventors have already demonstrated numeroushigh performance circuits and beam-formers at 2 GHz, 5 GHz, 15 GHz, 24GHz, and 26 GHz using 0.18 μm CMOS (TSMC) and SiGe BiCMOS (IBM7HP)processes. The new generation of SiGe HBT technology (IBM8HP) offered bythe IBM foundry have f_(max)=285 GHz, and is capable of generating morethan 10 dB power gain per transistor at the potential imaging frequencyof 94 GHz. Based on measurements at 60 and 77 GHz, these devices areexpected to achieve a noise figure (NF) of less than 6 dB at 94 GHz. Inaddition, this process includes 0.13 μm CMOS transistors, which allowsthe direct integration of imaging sensor's high speed digital signalprocessing (DSP) on the same wafer. Accordingly, in exemplaryembodiments of ring architectures, each amplifier can be a low-noisepower amplifier (LNNPA) to facilitate both transmit and receive caseswithout any switches as described before.

FIG. 10 is a photograph with circuit overlay of a 24 GHz 0.13 micronCMOS variable phase ring oscillator transmitter and receiverarchitecture 1000 according to an exemplary embodiment of the presentdisclosure. The architecture shown was constructed as a four-channel 24GHz CMOS phased array transceiver on a die approximately 2.15 mm by 2.35mm in size. A ring of tuned amplifiers 1002 is shown coupled to a phaseshifter 1004 and PLL 1006. Low noise amplifiers (LNA) 1008(1)-1008(4)are shown with power amplifiers 1010(1)-1010(4) as configured forcoupling to antenna (not shown).

FIG. 11 is an enlargement of a portion 1100 of FIG. 10 along with acorresponding circuit diagram showing. Diagram 1100 a shows a circuitdiagram of the ring of FIG. 10 with phase control. Picture 1100 b showsthe enlargement of area 1100 of FIG. 10.

Exemplary Embodiments Micro Radiators

Micro radiators useful as antenna according to the present disclosurecan be configured as on-chip differential radiating elements that act asthe resonant load of tuned amplifiers at each node of the proposed ringarchitecture. Formation and radiation of phased array pattern can beaccomplished in one step. From this point of view, the overall systemefficiency can be defined as the ratio of radiated electromagneticenergy in the desired direction to the energy drawn from the powersource. In order to maximize the system efficiency, each nonlinear blockof the ring configuration should convert most of the DC power to astored power in standing wave EM fields that are radiated. The small(sub wavelength) spacing between these radiators removes the unwantedgrating lobes out of the visible space for wide beam scanning. Unliketraditional implementations of phased arrays, the coupling betweenantenna elements is not a concern. The reason is that desired beampatterns are formed by setting EM boundary condition (amplitude andphase) on a silicon wafer via a closely spaced mesh of micro-radiators.Micro-radiators work collectively to set a boundary conditioncorresponding to a desired beam pattern.

Proper lengths of microstrips form a standing wave at the desiredfrequency that potentially couples to air for radiation. In a standardsilicon process, the proximity of the ground plane to the conductor(typical space between the topmost and bottommost metal layers is lessthan 15 μm) causes the image current to cancel the radiation of thesignal current. The extent of radiation is therefore proportional to thesignal-ground separation, limiting the radiation efficiency to less than1-2%. Radiation efficiency can be increased by placing the ground planeat the back side of silicon chip. Unfortunately, the electromagneticfields in this case penetrate the lossy silicon substrate where asignificant portion of energy is dissipated. In exemplary embodiments,the ratio efficiency of on-chip ring antennas may be maximize oroptimized by tapering techniques and also by using floating metalshields under the antenna. These floating metals reduce electricallength of the antenna (slow wave structure) and reduce the loss.

Exemplary embodiments of these radiators can be realized on the topmetal layer of a standard silicon process that is 15 μm above the 10Ω-cmsubstrate with a thickness of 250 μm. The radiation efficiency of theseantennas are 10%, and 15% at 90 GHz, respectively, which issignificantly higher compared to previously reported on-chip antennas[20][21]. We anticipate increasing the antenna efficiency to around 30%with our proposed tapering scheme and slow wave structure.

Exemplary Applications

Exemplary embodiments of the present disclosure can be used in twodifferent configurations: either as a standalone handheld device, or asa unit cell in a collaborative network of similar devices that arerandomly placed or tiled in the desired environment under controlembodiments of the present disclosure can serve under various scenariosfor wireless communication and imaging including the following:

Solider and Personnel Safety

Embodiments of the present disclosure can be placed as embedded sensorsin clothing and can be networked with all other personnel and vehicles.It can be configured as hand-held device (personnel low-range RADAR) forthe detection of enemy, improvised explosive devices (IED), and injuredsoldiers in search and rescue missions.

Cooperative Autonomous Agents

Embodiments of the present disclosure can be used as both an imagingsystem and a communication device in a network of cooperative autonomousagents in the battlefield such as unmanned tanks and unmanned aerialvehicles (UAV).

Multi-Function Wireless Communication

Each embodiments of the present disclosure silicon wafer can be used formulti-function wireless communication and imaging device where multipleelectromagnetic beams at different frequencies can be formedconcurrently, each beam for a particular application (e.g., independentbeams for voice, data, global positioning system, RADAR).

Exemplary embodiments may provide radar for automobile applications,e.g., collision avoidance/mitigation. It is expected that in the nearfuture all cars will be equipped with as many as 10 of such radars forspeed control, collision avoidance warning, blind spot detection,parking assistance, and airbag inflation assistance with the ultimategoal of having an intelligent transportation system.

Automotive radars at both 24 GHz and 77 GHz frequency bands have beenrecently deployed in the luxury cars such as Mercedes Benz, BMW, andLexus as an option. These radar systems, many of them not scanningarrays, use high cost compound semiconductor technologies andconventional architectures. Therefore, they cost a few thousand dollars,a price unacceptable for most mainstream vehicles.

Surveillance

Embodiments of the present disclosure wafers can be embedded in thebattlefield for distributed sensing embodiments of the presentdisclosure nodes can image the environment in a collaborative network.Each node can use its beam to image and/or to communicate theinformation to other nodes. Collaborative sensing is robust to nodefailures, reconfigurable, secure, power efficient, and achieves higherrange and resolution.

The application of embodiments of the present disclosure to any ofaforementioned scenarios requires low cost, low power consumption from asingle source of energy (e.g., solar cell or small battery), highspatial resolution for the scanned beam, and reliability. The choice offrequency(ies) and maximum output power derives from a tradeoff betweenthe required spatial resolution, communication data-rate, and devicesize in one hand and maximum range on the other hand. For instance, in a16×16 array where the center frequency is 94 GHz, the total chip area isless than 2.5 cm×2.5 cm and the spatial resolution at a distance of 10 mis roughly 1.5 m. In a wafer scale integration of the same system, anarray of 256×256 can be integrated resulting in a spatial resolution ofbetter than 8 cm at 10 m distance. The depth resolution in the imagingscenario and the bit-rate in the communication scenario are inverselyproportional and proportional to signal bandwidth, respectively.Assuming the maximum output power of 10 dBm for each array element inthe transmit mode and the element noise figure of 8 dB in the receivemode, instantaneous bandwidth of 100 MHz around 90 GHz, elementefficiency of 10%, and minimum received SNR of 10 dB at each element, a16×16 array can achieve the desired range of 10 m for RADAR and imagingapplications. If such a system is used for communication, a maximumdata-rate of 11.3 Mbs can be achieved assuming uncorrelated noise ateach element. Relatively speaking, to achieve the same beam width (RADARangular resolution) at 24 GHz, the chip dimensions has to increase byalmost a factor of 4, while the range improves by more than a factor of2. The communication bandwidth is directly proportional to thebandwidth.

Besides direct military applications, in future embodiments of thepresent disclosure can be transferred to serve in numerous commercialsettings including low-cost automotive RADAR, short-range high data-rate(Gbit/sec) WLAN, biomedical imaging, sensory networks in factories forquality-control or for environmental monitoring.

An exemplary embodiment of the present disclosure provides a 4-elementphased array chip that uses the variable phase ring oscillatorarchitecture fabricated in a 0.13μ CMOS technology that targets the 24GHz frequency band.

Accordingly, embodiments of the present disclosure can provide for novelfully integrated architectures and RF circuits to generate and receivearbitrary two-dimensional beam patterns at single frequency or multipleconcurrent frequencies. The central theme of this effort is therealization of an Active Silicon Programmable Integrated Radiator(embodiments of the present disclosure) embodiments of the presentdisclosure is a new class of radiator, communication device, or imagingsensor where the EM at the radiator aperture is actively controlled in amicron scale on a single silicon wafer, resulting in an unprecedentedfunctionality, flexibility, reliability, and cost reduction. Theamplitude and phase of the EM field at the aperture of the so-calledmicro-radiators will be controlled by active silicon devices, namelyComplementary Metal Oxide Semiconductor Field Effect Transistors (CMOSFET) or Silicon Germanium Hetero-structure Bipolar Junction Transistors(SiGe HBT).

In addition to the automotive radar application, our invention isapplicable to other high resolution ranging and imaging systems as wellas high data-rate wireless communications such as those suitable forwireless local area networks (WLAN).

While certain embodiments have been described herein, it will beunderstood by one skilled in the art that the methods, systems, andapparatus of the present disclosure may be embodied in other specificforms without departing from the spirit thereof. For example, whilearrays including a specific number of elements and/or spacing (e.g.,λ/2) have been shown and described, other configurations and embodimentsmay be used and are within the scope of the present disclosure.

Accordingly, the embodiments described herein are to be considered inall respects as illustrative of the present disclosure and notrestrictive.

1. A variable-phase ring-oscillator circuit comprising: a plurality oftuned amplifiers tuned amplifier configured as a ring in series; a firstphase shifter coupled to the plurality of tuned amplifiers tunedamplifier, wherein the first phase shifter provides a phase delaybetween each pair of successive tuned oscillators; and a phase-lockedloop connected to the ring.
 2. The variable-phase ring-oscillatorcircuit of claim 1, further comprising a second phase shifter connectedto the ring.
 3. The variable-phase ring-oscillator of claim 1, whereinthe phase-locked loop includes a frequency divider.
 4. Thevariable-phase ring-oscillator of claim 1, wherein the phase-locked loopincludes a frequency phase detector.
 5. The variable-phasering-oscillator of claim 1, wherein the phase-locked loop includes acharge pump.
 6. The variable-phase ring-oscillator of claim 1, whereinthe phase-locked loop includes a loop filter.
 7. The variable-phasering-oscillator circuit of claim 1, further comprising a plurality ofantennas coupled to the plurality of tuned amplifiers, wherein theplurality of antennas are configured and arranged as an array.
 8. Thevariable-phase ring-oscillator circuit of claim 1, wherein the pluralityof tunable amplifiers and plurality of antennas are configured on asemiconductor substrate.
 9. The variable-phase ring oscillator of claim2, wherein the phase-locked loop in disposed on a semiconductorsubstrate.
 10. The variable phase ring oscillator circuit of claim 1,wherein the plurality of tuned amplifiers are configured in a twodimensional m×n array.
 11. The variable phase ring oscillator circuit ofclaim 10, further comprising m first phase shifter, each coupled to aseparate ring in the m dimension of the m×n array and n second phaseshifter, each coupled to a separate ring in the m×n array.
 12. A phasedarray comprising: a plurality of tuned amplifiers configured in seriesin a ring; a first phase shifter inserted in the ring coupled to theplurality of tuned amplifiers, wherein the first phase delay structureprovides a phase delay between each pair of successive tuned amplifier;a phase-locked loop connected to the ring; and a plurality of antennasconnected to the ring.
 13. The phased array of claim 12, wherein theplurality of tuned amplifiers are disposed on a semiconductor substrate.14. The phased array of claim 13, wherein the substrate comprisesilicon.
 15. The phased array of claim 13, wherein transistors in thecircuit comprise bipolar, FET, or hetero-junction bipolar transistors(HBT).
 16. The phased array of claim 15, comprising CMOS, BJT, orsilicon-germanium transistors.
 17. The phased array of claim 12, whereinplurality of tuned amplifiers comprises a plurality of amplifierdifferential pairs.
 18. The phased array of claim 13, wherein thephase-locked loop is disposed on the substrate.
 19. The phased array ofclaim 18, wherein the plurality of antennas are disposed on thesubstrate.
 20. The phased array of claim 12, further comprising a secondphase shifter inserted in second ring coupled to the plurality of tunedamplifiers, wherein the second phase delay structure provides a secondphase delay between each pair of successive tuned amplifiers.
 21. Amethod of using a variable phase ring oscillator comprising: applying asignal to a variable phase ring oscillator coupled to a plurality ofantennas; providing a phase shift to the signal within the ringoscillator; locking the phase and/or frequency of the ring oscillatorwith a phase locked loop; and associating a phase shift and amplitude ofa signal with a direction for the plurality of antennas.
 22. The methodof claim 21, wherein applying a signal to a variable phase ringoscillator comprises transmitting a signal from variable phase ringoscillator coupled to the plurality of antennas.
 23. The method of claim21, wherein applying a signal to a variable phase ring oscillatorcomprises receiving a signal from variable phase ring oscillator coupledto the plurality of antennas.
 24. The method of claim 22, furthercomprising modulating a carrier signal.
 25. The method of claim 24,wherein modulating a carrier signal comprises a phase or frequencymodulation scheme.
 26. The method of claim 24, wherein the modulationscheme comprises FM, PM, PSK, FSK modulation.
 27. The method of claim23, further comprising demodulating a carrier signal.
 28. The method ofclaim 27, wherein demodulating a carrier signal comprises a phase,frequency, or demodulation scheme.
 29. The method of claim 24, whereinthe demodulation scheme is selected from the group consisting of QAM,OFDM, BPSK, AM, FM, PM, QPSK, FSK, and PSK.
 30. The method of claim 21,further comprising determining a distance to a target object using thesignal applied to the variable phase ring oscillator coupled to aplurality of antennas.
 31. The method of claim 21, wherein the signalapplied is a GHz signal.
 32. The method of claim 31, wherein the signalapplied is in a 2.4 GHz, 5 GHz, 22-29 GHz, 59-64 GHz, 71-76 GHz, 77-78GHz, 81-86 GHz, or 92-95 GHz band.
 33. The method of claim 30, whereindetermining a distance comprises determining a distance from an objectto an automobile connected to the variable phase ring oscillator coupledto the plurality of antennas.
 34. The method claim 33, furthercomprising determining relative direction from the automobile to theobject.